Direct-conversion is an alternative wireless transmitter architecture to the well-established superheterodyne, particularly for highly integrated, low-power terminals. Its fundamental advantage is that the transmitted signal is mixed directly to radio frequency (RF) without the use of an intermediate frequency. This means less hardware complexity, lower component count and a simpler frequency plan.
In direct-conversion transmitters the most serious problem is that of local oscillator (“LO”) feed-through due to a variety of on-chip coupling mechanisms. While the LO is necessary to convert the baseband signal to RF for transmission, if significant LO leakage occurs in the middle of the up-converted signal spectrum, the signal will be corrupted and rendered unreceivable.
Furthermore, in modern communication systems, the transmitter must be able to control the power that it is transmitting. The LO feed-through amplitude cannot be controlled as it is an unwanted byproduct of circuit non-idealities. This forces the transmitting signal to always be much larger than the LO feed-through.
Traditionally, LO feed-through can be reduced through the use of on-chip or external calibration schemes at the cost of additional hardware size and complexity.
Another problem in direct-conversion transmitters is in-phase and quadrature (“IQ”) imbalance of the LO and signal path. Traditionally, direct conversion transmitters and receivers need a local oscillator with quadrature outputs for vector modulation and demodulation. However, when the quadrature signal components are not equal in amplitude and not exactly 90 degrees out of phase, the signal degrades in quality.
Quadrature phases are typically derived by passing a reference local oscillator through a CR-RC phase shift network. Ideally, this creates two signals with equal amplitude and 90 degrees of phase difference. However, this depends on the accuracy of resistors and capacitors which make up the phase shift network. The resistors and capacitors can vary by up to 15 percent in a typical integrated circuit causing the in-phase and quadrature components to have different amplitudes and a phase difference not equal to 90 degrees.
In addition, layout differences between the in-phase and quadrature paths can cause additional amplitude/phase imbalance. Contributing to further in-phase/quadrature imbalance, the circuits in the in-phase and quadrature paths (i.e. amplifiers and mixers) have physical properties that differ. Many feedback calibration schemes have been proposed and implemented to mitigate quadrature imbalance at the cost of hardware and/or system complexity.
A conventional direct conversion transmitter is illustrated in FIG. 1. As illustrated in FIG. 1, a direct conversion transmitter takes the modulated in-phase and quadrature baseband signals 100, 101 and translates the signal to an RF frequency by multiplying them by a quadrature LO. Specifically, the transmitter multiplies the baseband signals 100, 101 with two different phases 102, 103 of a local oscillator 107 using a first mixer 110 and a second mixer 111. The in-phase and quadrature baseband signals can be denoted BBI and BBQ. The two phases 102, 103 of the local oscillator are 90 degrees apart and thus, are known as the in-phase (“I”) 102 and quadrature (“Q”) 103 components and can be denoted LOI and LOQ respectively. The mixer outputs 104, 105 are summed together to produce the RF signal 106. The RF signal 106 can be represented by Equation 1.RF=BBI×LOI+BBQ×LOQ  Equation 1
Another conventional direct conversion architecture is shown in FIG. 2. This differential direct conversion architecture is more resilient to self generated noise than the one illustrated in FIG. 1. In FIG. 2, the I and Q baseband signals are differential signals composed of positive and negative components. The equivalent single-ended baseband in-phase signal to the differential baseband in-phase signal is described by Equation 2. In Equation 2, BBI denotes the single-ended in-phase signal, BBI,pos denotes the positive component of the differential signal, and BBI,neg denotes the negative component of the differential signal. A differential signal has a positive and negative component which subtract to form the signal. Equation 3 describes the quadrature relationship in similar terms.BBI=(BBI,pos−BBI,neg)  Equation 2BBQ=(BBQ,pos−BBQ,neg)  Equation 3In Equation 2 and Equation 3, BBI,pos, BBI,neg, BBQ,pos, BBQ,neg correspond to 200, 201, 202, 203, respectively, in FIG. 2.
The differential direct conversion architecture shown in FIG. 2 uses differential LO signals to mix the baseband signal to RF. The polyphase network 205 is a circuit which converts the local oscillator voltage waveform 204 into four voltage waveforms 206, 207, 208, 209 at the same frequency as LO but at 0, 180, 90, 270 degrees offset compared to the local oscillator signal 204 respectively.
Collectively, these four signals 206, 207, 208, 209 are referred to as polyphase local oscillator signals. To facilitate the present description, the following signals 206, 207, 208, 209 are denoted as LO0, LO180, LO90, LO270 corresponding to their phase shift compared to the local oscillator 204. Shifting a sinusoidal signal 180 degrees in phase is the same as inverting the signal. Therefore, the equivalent single-ended in-phase and quadrature LO signals, denoted LOI and LOQ, are described mathematically in Equations 4 and 5.LOI=LO0−LO180  Equation 4LOQ=LO90−LO270  Equation 5The differential quadrature baseband signals are routed to the differential mixers 210, 211 where they are multiplied by the differential local oscillator signals.
At the first mixer 210, the differential in-phase baseband signal (BBI) is multiplied by the in-phase LO (LOI). Likewise, at the second mixer 211, the differential quadrature baseband signal (BBQ) is multiplied by the quadrature LO (LOQ). Equation 6 describes the mixing process and summing process 212 of the differential signals to generate the RF signal 213. Likewise, Equation 7 is a simplified version of Equation 6 where all the differential signals are represented as single-ended signals.RF=(BBI,pos−BBI,neg)×(LO0−LO180)+(BBQ,pos−BBQ,neg)×(LO90−LO270)  Equation 6RF=BBI×LOI+BBQ×LOQ  Equation 7
Now, to elucidate the problems with direct conversion transmitters, LO feed-through and imbalance distortions will be added to Equations 6 and 7. LO feed-through is added to the output of the mixers. LOC1 represents the differential LO feed-through from the polyphase LO signal to the output of mixer 210, and LOC2 represents the differential LO feed-through at the at output of mixer 211. Likewise the amplitude and phase imbalance of the mixers and the polyphase LO signals can be accounted for at the output of each mixer. A complex multiplicative term, A1ejP1, represents a random amplitude variation (“A1”) and a random phase variation (“P1”) introduced by the first mixer 210, the signal path 200, 201 and LO path 206, 207 connected to the mixer. Likewise, A2eP2 represents a random amplitude and phase variation introduced by the second mixer 211 and the signal 202, 203 and LO paths 208, 209 connected to it. Thus, with these distortions added, Equations 6 and 7 become Equations 8 and 9.RF=(BBI,pos−BBI,neg)×(LO0−LO180)×A1ejP1=LOC1+(BBQ,pos−BBQ,neg)×(LO90−LO270)×A2ejP2=LOC2  8RF=BBI×LOI×A1ejP1=LOC1=BBQ×LOQ×A2ejP2=LOC2  Equation 9
As seen in Equations 8 and 9, the RF distortion grows as A1 and A2 differ and as P1 and P2 differ. As the distortion increase, it is harder for the signal to be received and decoded. Likewise as LOC1 and LOC2 increase, they become the dominant component of the RF signal in Equation 8 and 9.
There is thus a need for a transmitter for overcoming these problems and/or providing general improvements over prior art transmitters.